Balanced processing based on receiver selection

ABSTRACT

A device and method for receiving a radio signal transmitted over a channel, selecting and deselecting an equalizer, and balancing processing in response to such selection. A radio frequency signal is received from a transmission channel into a receive path, and delay spread in the radio frequency signal is estimated using an analysis circuit. The analysis circuit also determines a threshold delay spread. In the event the estimated delay spread exceeds the threshold delay spread, an equalizer is selected, otherwise the equalizer is deselected. Similarly, in the event the estimated delay spread does not exceed the threshold delay spread, a high complexity processor is selected, otherwise a low complexity processor is selected. If the low complexity processor is selected, an output signal is generated using the low complexity processor, and if the high complexity processor is selected, the output signal is generated using the high complexity processor.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation-in-part of Applications Ser. No.08/273,661, filed Jul. 11, 1994, entitled "Time-Reversed InfiniteImpulse Response Filtering of an Asymmetric Signal", now allowed, andSer. No. 08/246,851, filed May 19, 1994, now pending, entitled "ReceiverSelection Based on Delay Spread Estimation" by Michael Parr, and LongHuynh and Michael Parr, respectively, and assigned to the assignee ofthe present application, the priorities of which are hereby claimed andthe disclosures of which are hereby incorporated by reference fullyherein.

BACKGROUND OF THE INVENTION

The present invention relates to balanced processing based on selectionof a receiver, and more particularly to delay spread estimation for aradio frequency signal path, selection of a receiver equalizer based onsuch estimation, and balanced processing based on such selection. Evenmore particularly, the present invention relates to delay spreadestimation of a cellular telephone signal path, selection of a receiverequalizer based on such estimation, wherein a delay spread threshold forselection of the receiver equalizer is a function of signal-to-noiseratio, and balanced voice processing based on such receiver selection.

Communication channels in the cellular environment commonly impose acombination of distorting effects on transmitted signals. Rayleighfading, where a signal's perceived power level rises and falls rapidlyover a wide range, results from the combination (interference) ofsignals that have traversed paths differing in length by at least asignificant fraction of a wavelength (i.e., about 30 cm. for cellular).This type of interference is known as multi-path interference.Differences in path transmission times that approach the time taken totransmit a symbol result in a second problem called delay spread.

Delay spread results in reception of multiple delayed replicas of atransmitted signal. Each Rayleigh faded replica has randomly distributedamplitude and phase, and the rate at which this complex quantity variesis constrained by the Doppler bandwidth associated with a vehicle'sspeed. In a frequency nonselective environment, the sampled outputs of areceiver's matched filter provide uncorrelated estimates of thetransmitted data. As such, in terms of discrete time samples, thechannel has exhibited an impulse response proportional to a deltafunction. With delay spread, on the other hand, the discrete timechannel impulse response is extended to introduce energy at a number ofsymbol times. The effect of the channel on the transmitted signal, inturn, may be viewed as the convolution of the transmitted informationwith the channel's impulse response. The channel, therefore, emulates aconvolutional coding process (encoder).

This leads to the possibility of estimating the transmitted informationthrough the use of methods analogous to typical decoding ofconvolutional codes, i.e., maximum likelihood sequence estimationtechniques.

Such maximum likelihood sequence estimation techniques, when implementedin a cellular telephone receiver, provide improved performance in thereceive path, when multiple replicas of the transmitted signals arereceived. However, when multiple replicas are not received, i.e., whenthe transmitted signal is received having traversed a single signalpath, such maximum likelihood sequence estimation receivers actuallydegrade performance. Thus, it is desirable to switch between a receiverthat performs maximum likelihood sequence estimation, and a receiverthat does not perform such maximum likelihood sequence estimation.

Both of these receivers can be implemented using a digital signalprocessor. Thus it is desirable to load and execute either an equalizingreceiver routine or a non-equalizing receiver routine in the digitalsignal processor depending on whether multi-path interference isdetected. The equalizing receiver requires more processing from thedigital signal processor.

For the North American digital cellular system, a number of documentsdefine the standards of implemented components. With respect to thisinvention, the following are of interest: "Dual-Mode Mobile Station-BaseStation Compatibility Standard" denoted here as IS-54, EIA/TIA ProjectNumber 2398, Rev. A Jan 1991; "Recommended Minimum Performance Standardsfor 800 MHz Dual-Mode Mobile Stations", denoted here as IS-55, EIA/TIAProject Number 2216, Apr. 1991; and Recommended Minimum PerformanceStandards for Full Rate Speech Codes denoted here as IS-85, TIA/EIA, May1992.

In accordance with these standards, a voice encoder/decoder is used toperform voice processing, or speech coding. Voice processing is not,however, specified by the above specifications with bit-exactdescription. Instead such voice processing is described in functionalform, leaving the exact implementation of the voice processing to thedesigner. Thus, more complex voice processing techniques can be utilizedto achieve higher quality voice output, and lower quality voiceprocessing techniques can be utilized to achieve lower quality voiceoutput, while still conforming to the above-mentioned standards. Thehigher quality voice processing techniques employ more complicatedprocessing, and therefore require more powerful and more costly, oradditional digital signal processors to implement. This is particularlytrue if the equalizing receiver, which also employs more complicatedprocessing, is to be implemented simultaneously with such higher qualityvoice processing techniques.

The present invention advantageously provides for balanced processing,which allows for the use of higher quality voice processing (or otherhigh complexity processing), and selection between equalizing andnon-equalizing receivers, while minimizing the digital signal processingpower needed. As a result, the invention also minimizes the costrequired, for implementation.

SUMMARY OF THE INVENTION

The present invention advantageously provides for balanced processingwithin a digital signal processor based on selection of a receiver.

The present invention can be characterized as a method of selecting anddeselecting an equalizer, and of balancing processing in response tosuch selection. The method includes (a) receiving a radio frequencysignal from a transmission channel into a receive path, and (b)estimating a preselected characteristic of the transmission channel,e.g., an estimated delay spread in the radio frequency signal. Themethod also includes (c) determining a preselected threshold, e.g., athreshold delay spread, and (d) selecting the equalizer in the event theestimated delay spread exceeds the threshold delay spread, or (e)deselecting the equalizer in the event the estimated delay spread doesnot exceed the threshold delay spread. In addition, the method includes(f) selecting a high complexity processor in the event the estimateddelay spread does not exceed the threshold delay spread, and (g)selecting a low complexity processor in the event the estimated delayspread exceeds the threshold delay spread. Finally, the method includes(h) generating a transmitter output signal using the low complexityprocessor in the event the low complexity processor is selected, or (i)generating the transmitter output signal using the high complexityprocessor in the event the high complexity processor is selected. Whenselected, the equalizer receives the radio frequency signal from thereceive path, and generates the equalizer signal in response thereto.

The invention can also be characterized as a communications device forreceiving a radio frequency signal transmitted over a channel. Thecommunication device includes a radio receiver that receives the radiofrequency signal from a transmission channel into a receive path, and ananalysis circuit coupled to the radio receiver. The analysis circuitestimates a preselected characteristic of the transmission channel,e.g., an estimated delay spread in the radio signal, and determines apreselected threshold, e.g., a threshold delay spread. The device alsoincludes an equalizer that equalizes the radio frequency signal andgenerates an equalizer signal in response thereto. The equalizer iscoupled to the radio receiver. A first switching device is coupled tothe analysis circuit, the equalizer and the radio receiver. The firstswitching device selects the equalizer in response to the analysiscircuit in the event the estimated delay spread exceeds the thresholddelay spread, and deselects the equalizer in the event the estimateddelay spread does not exceed the threshold delay spread.

The device further includes a radio transmitter that transmits atransmitter signal from a transmission path into the transmissionchannel, a high complexity processor coupled to the radio transmitter, alow complexity processor coupled to the radio transmitter, and a secondswitching device coupled to the high complexity processor and the lowcomplexity processor. The second switching device receives a basebandsignal that is coupled by the second switching device to the highcomplexity processor in the event the estimated delay spread does notexceed the threshold delay spread, and is coupled to the low complexityprocessor in the event the estimated delay spread exceeds the thresholddelay spread.

BRIEF DESCRIPTION OF THE DRAWINGS

The above and other aspects, features and advantages of the presentinvention will be more apparent from the following more particulardescription thereof, presented in conjunction with the followingdrawings wherein:

FIGS. 1 and 1A-1C are block diagrams of an embodiment of an apparatusmade in accordance with the present invention;

FIG. 2 is a flow chart of the steps traversed by a maximum likelihoodsequence estimation equalizing receiver that can be used with theembodiment of FIG. 1;

FIG. 3 is a flow chart of the steps traversed by the maximum likelihoodsequence estimation equalizing receiver of FIG. 2 in order to implementcarrier frequency offset compensation;

FIG. 4A and 4B flow charts of the steps traversed by the maximumlikelihood sequence estimation equalizing receiver of FIG. 2 in order toimplement bit timing; and

FIG. 5 are graphs showing an exemplary threshold delay spread used inthe embodiment of FIG. 1 to determine whether the maximum likelihoodsequence estimation equalizing receiver or a non-equalizing receivershould be utilized.

Corresponding reference characters indicate corresponding componentsthroughout the several views of the drawings.

DETAILED DESCRIPTION OF THE INVENTION

The following description of the presently contemplated best mode ofpracticing the invention is not to be taken in a limiting sense, but ismade merely for the purpose of describing the general principles of theinvention. The scope of the invention should be determined withreference to the claims.

Referring first FIGS. 1A-1C, a block diagram is shown of an embodimentof an apparatus made in accordance with the present invention. A digitalcellular mobile telephone transceiver system 20 incorporates a receivechannel path 14 and a transmit channel path 16. The receive channel path14 employs a maximum likelihood sequence estimation equalizer 21 (orequalizing receiver), a non-equalizing detector system 18 (ornon-equalizing receiver), and an equalizer activation system 19. Thereceive channel path 14 comprises an amplifier 22 whose output iscoupled by way of a down-converter (or demodulator), comprising afrequency source 23 and a mixer 24, to an analog filter 25. An analog todigital converter 26 is coupled to the analog filter 25 in order todigitize the down-converted data. A matched filter 27 is coupled betweenthe analog to digital converter 26, and a memory 30 (or buffer). Theequalizer 21 is coupled to the memory 30 via a switching device 41. Thememory 30 is coupled to a common pole of the switching device 41 and theequalizer 21 is coupled to a second pole of the switching device 41. Theequalizer 21 comprises a 4-state equalization trellis 31 that is adaptedto calculate maximum likelihood sequence estimation metrics, a channelimpulse response estimator 32, and an equalizer control circuit 33. Theequalization trellis 31 is coupled to the memory 30 and is also coupledto a switching device 49 at a second pole. The channel impulse responseestimator 32 is coupled to the memory 30 via the switching device 41 atits second pole, and is coupled to the switching device 49 at its secondpole. The channel impulse response estimator 32 is also coupled to theequalization trellis 31, and to the equalizer control circuit 33. Boththe equalizing receiver 21 and the non-equalizing receiver 18 canpreferably be implemented as software routines executed on a digitalsignal processor (not shown). The equalizing receiver 21, whenimplemented as a software routine, is more complex, and therefore hasgreater processing requirements than the non-equalizing receiver 18,when implemented as a software routine.

Also coupled to the memory 30 via the switch 41 is the non-equalizingdetector system 18. The non-equalizing detector system 18 is coupled tothe switching device 41 at a first pole of the switching device 41, andcomprises a non-equalizing detector 43 and a control unit 45. Suitablenon-equalizing detector systems 18 are known to those skilled in theart. Several such non-equalizing detector systems 18 are shown in thereference "Digital Communications 2nd Edition", by J. G. Proakis, at,e.g., Chapter 4, 1989. The non-equalizing detector 43 is coupled to thememory 30 via the switching device 41 at its first pole, and is coupledto a receiver output 47 via another switching device 49 at its firstpole. The receiver output 47 is coupled to a common pole of theswitching device 49. The control unit 45 is coupled to thenon-equalizing detector 43, and generates bit timing control signals andfrequency control signals analogous to those generated by the equalizercontrol circuit 33, described hereinbelow.

The control unit 45 is coupled to the frequency source 23 via aswitching device 48 and is coupled to bit timing control circuitry 37via another switching device 46. The control unit 45 is coupled to theswitching device 48 at a first pole, and the frequency source 23 iscoupled to the switching device 48 at a common pole. Similarly, thecontrol unit 45 is coupled to the switching device 46 at a first pole,and the bit timing control circuitry 37 is coupled to the switchingdevice 46 at a common pole.

A serially coupled AGC circuit 35 and gain control circuit 38 arecoupled to the amplifier 22. The equalizer control circuit 33 is coupledto an output of the analog to digital converter 26 via time offsetmatched filters 34 and is coupled to an input to the frequency source 23via a second pole of the switching device 48. Symbol sampling (bittiming) time control circuitry 37 is coupled to the equalizer controlcircuit 33 via a second pole of the switching device 46 and provides bittiming control signals to the analog to digital converter 26. The bittiming control circuitry 37 is also coupled to the acquisition circuit36. The output of the matched filters 27 is coupled to the AGC circuit35 and the acquisition circuit 36 and to the equalizer control circuit33 that is employed to control the frequency source 23 and providetraining data for use in initializing the equalizer 21.

The transmit channel path 16 comprises a switching device 28, a highcomplexity voice processor 29, a low complexity voice processor 39, amultiplier 44 and a frequency source. A baseband data signal generatedby commonly known transmitter circuits (not shown) within the cellulartransceiver 20 is received into a common pole of the switching device28. A first pole of the switching device 28 is coupled to the highcomplexity voice processor 29, and a second pole of the switching device28 is coupled to the low complexity voice processor 39. The highcomplexity voice processor 29 provides superior quality voice output,i.e., superior in clarity and accuracy, but requires processing overheadin excess of that which is required by the low complexity voiceprocessor 39. For example, the low complexity voice processor 39 canutilize only a portion, e.g., every other one, of the set of codebookentries specified by IS-54B and IS-85, thereby reducing the codebooklookup processing requirements of the low complexity voice processor 39to half of what they would normally be. Furthermore, the low complexityvoice processor 39 can perform other calculations specified by IS-54Band IS-85 to a lesser degree of accuracy than the high complexity voiceprocessor 58. Each of these features of the low complexity voiceprocessor 39 results in reduced processing overhead and reduced qualityvoice output.

Both the low complexity voice processor 39 and the high complexity voiceprocessor 29 are preferably implemented as software routines thatexecute on the same digital signal processor (not shown) as is used toexecute the software routines that implement the equalizing receiver 21and the non-equalizing receiver 18.

Both the high complexity voice processor 29 and the low complexity voiceprocessor 39 are coupled to and input of the multiplier 44 (ormultiplexer). Another input of the multiplexer 44 is coupled to thefrequency source 50, which generates a modulation frequency (or carrierfrequency). Together the multiplexer 44 and frequency source 50 serve asa modulator, such as are known in the art, which modulates the output ofthe high and low complexity voice processors 29, 39 and prepares suchoutput for transmission.

The equalizer activation system 19 includes a ratio/SNR estimator 40 anda threshold decision controller 42. The threshold decision controller 42is coupled to the switching devices 41, 49, 48, 46, 28 and passes aswitch control signal thereto. The switching devices 41, 49, 48, 46, 28assume a first state or a second state in response to the switch controlsignal. In the first state, the switching devices 41, 49, 48, 46, 28couple their respective common poles to their respective first poles,and in the second state they couple their respective common poles totheir respective second poles.

When the switching devices 41, 49, 48, 46 are in the first state, thereceiver output 47 is generated by the non-equalizing detector system18, and when the switching devices 41, 49, 48, 46 are in the secondstate the receiver output 47 is generated by the equalizer 21.

When the switching device 28 is in the first state, the baseband datasignal from the transmitter circuits is processed by the high complexityvoice processor 29. When the switching device 28 is in the second state,the baseband data signal is processed by the low complexity voiceprocessor 39. Thus, the high complexity voice processor 29 is used onlyin combination with the non-equalizing receiver 18 and the lowcomplexity voice processor 39 is used only in combination with theequalizing receiver 21.

In this way, the higher processing demands of the equalizing receiver 21are balanced with the lower processing requirements of the lowcomplexity voice processor 39. Similarly, the lower processing demandsof the non-equalizing receiver 18 are balanced with the higherprocessing requirements of the high complexity voice processor 29.

In about 90% of environments in which cellular transceivers areutilized, there will be little or no multipath interference, andtherefore the high complexity voice processor 29/non-equalizing receiver18 combination will be utilized to produce superior performance, i.e.,clearer and cleaner voice output at the cellular transceiver and basestation. In the approximately 10% of cases where the equalizing receiver21 is required, the higher processing demands of the equalizing receiver21 are accommodated by switching to the low complexity voice processor39. As a result, the processing requirements within the cellulartransceiver 20 remain relatively constant to those of the highcomplexity voice processor 29/non-equalizing receiver 18 combination.Thus, processing power within the digital signal processor isefficiently utilized, and system cost can be held to a minimum, i.e.,the need for two digital signal processors and/or more powerful digitalsignal processors is eliminated.

In operation, a partially filtered RF signal with a center frequency of85.05 MHz enters the gain controllable amplifier 22. The resultingsignal is then down-converted using the frequency source 23 and themixer 24 to 461.7 kHz. This signal is then filtered using a narrowanalog filter 25 to reject most of the received signals outside the 30kHz band of interest. The resulting signal is then sampled and convertedto 8-bit digital samples using the analog to digital (A/D) converter 26.A 16-tap fractionally spaced digital FIR filter 27 then performs matchedfiltering to produce symbol spaced samples which enter the memory 30.Temporally offset matched filters 34 that are substantially the same asthe matched filters 27 are provided for use by the symbol timing controlcircuit 37, via the equalizer control circuit 33.

The principles of maximum likelihood sequence estimation employed in theequalizer 21 have been described in technical literature starting in theearly 1970's. A useful outline is found in "Adaptive Maximum-LikelihoodReceiver for carrier-Modulated Data Transmission Systems", by G.Ungerboeck, IEEE Tran. on Communications, Vol. COM-22, at, e.g., pp.624-636, May 1974. Another description of the maximum likelihoodsequence estimation technique is provided in the reference "DigitalCommunications 2nd Edition", by J. G. Proakis, at, e.g., pp. 610-642,1989.

The embodiment described herein makes use of an adaptation of theMaximum Likelihood Sequence Estimation (MLSE) equalizer described inU.S. Pat. No. 5,263,026, incorporated herein by reference. Variousaspects of the '026 patent MLSE equalizer are outlined and explainedherein with respect to the particular adaptation of the '026 patentutilized to implement a preferred embodiment of the present invention.

The maximum likelihood sequence estimation process is outlined asfollows. The channel has an impulse response containing significantenergy in, say, N symbols. Assume that the transmitter sends a sequenceof symbols, much longer than N. The transmitted sequence may bedescribed as the transitions between states, where each statecorresponds to a group of N-1 transmitted symbols. The states,therefore, correspond to overlapping groups of transmitted symbols. Inconsecutive states, therefore, all but one constituent symbol are thesame, and the possible transitions between states are correspondinglyconstrained. As each sample is received, the equalization trellis 31considers every possible sequence of N symbols that could havecontributed to its value, by convolving that sequence with the estimatechannel impulse response. For each hypothesized sequence, the result ofthe convolution corresponds, or fails to correspond, in some way(defined by a statistic called a metric) to the measured sample. On anindividual basis, the hypothesized sequence with the closest match tothe measured sample (the best metric) is the most likely to have beentransmitted. However, over many samples and under the constraint thatonly certain state transitions are possible, the path (sequence ofstates) with the minimum cumulative metric has maximum likelihood, andthis is what the decoder selects.

The system 20 has no a priori knowledge of the form of the encodercharacterizing the transmission channel. Performance of the equalizer 21therefore depends on the accuracy of the estimate of the encoder'sstate, the channel impulse response (CIR). The objective is to estimatethe form of the transversal finite impulse response filter that wouldtake as its input the transmitted information symbols {a(a)}, andproduce at its output the samples taken from the matched filter, {z(n)}.During the transmission of preambles and coded digital verificationcolor codes, the receiver knows the values of {a(n)}. However, at othertimes, only the estimated values {a_(d) (n)} are available for use inthe channel impulse response estimation process. The dependence leads toa significant performance-degrading possibility. If decision errorsemerge from the equalizer 21, and these are then used to update theestimate of the channel impulse response, then further decision errorsbecome more probable leading in a circular fashion to further decisionerrors and breakdown of the equalization process. This phenomena isreferred to as a "channel impulse response tracking breakdown". Suchdifficulties are most likely to arise at the periods of minimumsignal-to-noise ratio, or when the received signal power is at itsminimum during reception of a slot.

Within the IS-54 standard, which describes the interface between mobileand base equipment for North American digital cellular systems, eachinformation time slot is preceded by a known sequence, designated as thepreamble. As viewed by the receiver, therefore, information in the timeslot is bounded on both sides by known sequences; the preamble for thisslot and the preamble for the subsequent slot. Consequently, thisequalizer 21 is adapted to mitigate the effects of a channel impulseresponse tracking breakdown. By finding the most probable instant atwhich the problem might occur, equalizer 21 operation approaches thatinstant from both forward and a time-reversed directions, both of whichbegin with known information sequences that are useful for training.Assuming that a channel impulse response tracking breakdown occurs, thisapproach minimizes the number of affected symbols by predicting thefailure point and avoiding equalization beyond that point.

At 100 km/hr, which is the maximum speed specified in IS-55, whichdescribes the mobile unit minimum performance requirements, the averagetime between fades is on the order of 12 milliseconds. Given time slotdurations of about 6.7 milliseconds, there is only a small possibilityof two significant fades occurring within a time slot. However, veryclose to the center of the slot is the coded digital verification colorcode field. Even after a channel impulse response tracking breakdown,the channel impulse response estimator 32 is very likely to recoverduring processing of the coded digital verification color codes due tothe certainty of the transmitted date. Hence, the underlying period forwhich multiple fades are a concern is around 3.5 milliseconds. Thechance of more than one deep fade occurring during this time is verylow. Consequently, time-reversed equalization improves bit error rateperformance in the digital cellular environment.

The equalizer 21 uses a 4-state architecture, corresponding to N=2,where N is the length of the estimated channel impulse response. Thischoice assumes that the energy in two (symbol-spaced) samples of thechannel's impulse response dominates, To avoid channel impulse responsetracking breakdown problems, reverse equalization is used for thosesymbols following the minimum power point in a received time slot.

More specifically, FIG. 2 shows the processing performed in the maximumlikelihood sequence estimation based equalizer 21 of FIG. 1B. The firststep involves finding the location of the power fade (box 51) in termsof symbol number. Processing starts in the forward direction toward thelocation of the power fade. The symbol number is set to 0 (box 52), andthen incremented (box 53). A decision is made whether the symbol thenprocessed is a training symbol (box 54). If the symbol encountered is atraining symbol, then training data is inserted (box 57). If a trainingsymbol is not processed, then the equalization trellis is employed togenerate metrics and, if possible, a decision (box 55). This isaccomplished using equations outlined below. Then it is determined if adecision has been made (box 56). If a decision has been made, then anestimate of the channel impulse response is generated (box 58). If thedecision is not made, or once the channel impulse response estimate hasbeen generated, then the symbol number is compared to the location ofthe power fade plus a predetermined number of additional symbols (box59). Processing is then repeated by incrementing the symbol number (box53) and repeating steps (boxes 54-59) until the fade location plus apredetermined number of additional symbols has been reached.

Once the desired symbol location is reached in (box 59), then processingis performed in the reverse direction starting with the preamble of thenext succeeding time slot, namely symbol number 177, for example. Thesymbol number is set to 178 (box 62), and then decremented (Box 63). Adecision is made whether the symbol then processed is a training symbol(box 64). If the symbol encountered is a training symbol, then trainingdata is inserted (box 67). If a training symbol is not processed, thenthe equalization trellis is employed to generate branch metrics and adecision (box 65). This is accomplished using the equations outlinedbelow. Then it is determined if a decision has been made (ox 66). If adecision has been made then an estimate of the channel impulse responseis generated (box 68). If the decision is not made, or once the channelimpulse response estimate has been generated, then the symbol number iscompared to the location of the power fade less a predetermined numberof additional symbols (box 69). Processing is then repeated bydecrementing the symbol number (box 63) and repeating steps (boxes64-69) until the fade location less a predetermined number of additionalsymbols has been reached.

More particularly, and in operation, samples entering the equalizer 21may be identified as z(n), and the output decisions may be identified asa(n). The probability of correctness of a(n) is known with certainty thevalues of a(n), denoted a(n), are used by the channel impulse responseestimator 32 for training. At other times, the best estimate of a(n) isthe output of the traceback decision process of the equalization trellis31, denoted ad(n).

The equalization trellis 31 operates as follows. Equalization proceedsin the forward direction from the beginning of the preamble up until Msymbols after the minimum power symbol. In the reverse direction, thesame occurs with processing continuing M symbols beyond the minimumpower point. This overlap ensures that trace-back through the trellis inall likelihood converges to a single path by the minimum power point.

Traceback for actual decisions does not occur until the completion ofthe equalization process. In addition to final traceback, however, thereis a need for tentative decisions during equalization, to provide dataestimates for the channel impulse response estimation to remain current.A trade-off in determining these tentative decisions arises (a) becausethe more up-to-date the information is, the more up-to-date the channelimpulse response estimate can be (remembering that the channel is farfrom stationary at high speeds), and (b) the higher the number ofsymbols that are considered before tentative decisions are made, themore accurate the decisions will be and hence, the lower the probabilitythat errors are introduced into the channel impulse response estimation.In the case or 4-state equalization there is very little sensitivity tothe number of constraining lengths of delay introduced.

Branch metrics are calculated in the equalizer 21 using the followingequation: ##EQU1## where app₋₋ state (l) represents a hypothetical statein combination with potential input data; a_(h) (1,n) is correspondingtransmitted signal (constellation point), C represents the currentestimate of the channel's impulse response, and z is the measured outputof the matched filter 27.

The channel estimator 32 utilizes a second order least mean squarealgorithm to determine the coefficients of the transversal filter 27that is an estimate of the channel. ##EQU2## where C₀ (k) and C₁ (k) arecomplex values of estimated channel impulse response taps, C_(S0) (k)and C_(n) (k) are complex intermediate values related to the estimatedchannel impulse response taps, permitting second order operation, K₁ andK₂ within these equations control the rate of adaptation, and(conversely) the sensitivity to noise and decision errors. Consequently,to minimize the error rate, a trade-off between ability to track changesthe channel and degradation in performance due to imperfect inputinformation is needed to optimize the values of K₁ and K₂. The optimalvalues of K₁ and K₂ vary as a function of instantaneous signal to noiseratios and thus as a function of depth of fade. Therefore, algorithmsfor modifying the values during each burst have been evaluated, withconsiderable improvement in performance relative to that achievable withconstant settings.

One approach for modifying K₁ and K₂ provides good performance and is asfollows:

1. Set the values of K₁ and K₂ that will apply at the symbol determinedto correspond to the deepest fade; K₁₋₋ fade.

2. Adjust each value linearly (with preset slope--K₁₋₋ slope and K₂₋₋slope) to reach the selected values at the fade location, using:

before forward processing--initialize

    K.sub.1 =K.sub.1-- fade-K.sub.1-- slope.fade.sub.-- location

    K.sub.2 =K.sub.2-- fade-K.sub.2-- slope.fade.sub.-- location

before forward processing--initialize

    K.sub.1 =K.sub.1-- fade-K.sub.1-- slope.(177-fade.sub.-- location)

    K.sub.2 =K.sub.2-- fade-K.sub.2-- slope.(177-fade.sub.-- location)

during processing--as each symbol is processed

    K.sub.1 =K.sub.1 +K.sub.1-- slope

    K.sub.2 =K.sub.2 +K.sub.2- slope

where K₁₋₋ fade is the real value of K₁ at the symbol with the maximumestimated fade depth, K₂₋₋ fade is the real value of K₂ at the symbol ofmaximum estimated fade depth, K₁₋₋ slope is the real increment in K₁applied during processing of each symbol, K₂₋₋ slope is the realincrement in K₂ applied during processing of each symbol, and fadelocation is the symbol number at the maximum estimated safe depth, andlast₋₋ location is the symbol number of the final symbol.

Estimation of the location of the power fade entails use of the receivedsymbols from the matched filter 27, and the settings on the AGC circuit35 that were active during reception of those symbols., As the responseof the amplifier 22 to the AGC circuit settings is effectivelyinstantaneous, the primary delays in utilizing this information arise inthe matched filter 27. This filter 27 is a linear phase filter (constantdelay), so that available input information can be easily transformedinto an accurate estimate of the envelope power. This envelope isaveraged by a rectangular FIR filter over about ten symbol times, withvery food performance.

After completion of acquisition, the carrier frequency offset should beless than 200 Hz. To operate without impairment, this offset should beon the order of 20 Hz or less. Thus, estimation of and correction forcarrier offset must continue after acquisition. The method employedutilizes the fact that when frequency offset occurs, the taps of thechannel impulse response will rotate consistently at a rate proportionalto the offset. Changes in tap phases over fixed periods, therefore,provide an observable characteristic to apply to frequency control. Notethat random phase changes, so filtering is used to extract the frequencyoffset. In practice, offsets of around 1000 Hz can be resolved althoughthe maximum expected offset after acquisition is 200 Hz. The approachused is as follows:

1. During the reception of each burst, the half of that burst that doesnot include the deepest fade is selected for tracking. This scheme isaimed at avoidance of the very high rates of change in phase thattypically accompany transitions through low signal amplitudes.

2. Two samples of each of the two estimated channel impulse responsetaps are recorded: just after the preamble (or leading into thepostamble if the fade occurred during the first half of the slot), and20 symbols later (or 20 symbols earlier). At a symbol rate of 24,300symbols per second, a 100 Hz offset would result in an average rotationof 29.6 degrees during the 20 symbol period. For any rotation in excessof 180 degrees, the observed rotation would be less than 170 degrees butin the opposite direction. This aliasing could impact performance offrequency offsets above about 300 Hz. In typical operation, however, thedetriment to performance resulting from such aliasing has provedminimal, due to the anti-aliasing filtering inherent in the tracking.The selection of a sampling window of 20 symbols was based on concernabout this aliasing. Otherwise, a longer window would improve noiseimmunity.

3. From information determined during the bit timing fine tuning, thedominant tap is selected. Using the recorded settings for this tap, aphase change is calculated, yielding an estimate of the frequencyoffset.

4. These estimates are then filtered over many bursts to reduce the"noise" that arises primarily due to the random (zero mean) presence ofDoppler offsets and Gaussian noise. The filter output provides anestimate of the carrier offset and can be used to directly update thefrequency control hardware. The offset is given by:

    f.sub.-- offset.sub.-- estimate.sub.k +1=(1-K.sub.fo f.sub.-- offset.sub.-- estimate.sub.k +K.sub.fo freq.sub.-- observed

where freq₋₋ observed is derived from the observed phase change, theconstant K_(fo) controls the convergence rate of the estimation process,f₋₋ offset₋₋ estimate_(k) is the estimated frequency offset at frame"k", and K_(fo) is a constant controlling the convergence rate of thefrequency tracking. F₋₋ offset₋₋ estimate reaches half the resolution ofthe frequency source, then a step in frequency is applied, e.g., if theresolution is 20 Hz and f₋₋ offset₋₋ estimate exceeds 10 Hz, then a 20Hz change in reference is applied. At the same time f₋₋ offset₋₋estimate is reinitialized.

Referring to FIG. 3, a flow diagram is shown of the processing performedby the equalizer 21 to implement carrier frequency offset compensation.Utilizing an already located fade, a decision (box 100) is made as towhether to use the first or second half of the received slot forfrequency offset estimation. Based on this decision, samples are takentwenty symbols apart in the appropriate half of the slot (boxes 101,102). For the selected case, individual taps are compared and the largeris chosen (decisions 103, 104). The phases of the chosen tap at theselected two times are then subtracted (boxes 105-108) to produce"freq₋₋ observed", a noisy estimate of the offset. This is filtered (box109) to generate an accurate estimate of the offset. If an adjustment insetting of the frequency control would reduce this offset, then adecision is made to do so (decision 110); and the decision is thenimplemented (box 111).

The equalizer 21 is reasonably insensitive to errors in bit timing.However, for the following reasons, symbol timing adjustments continueduring equalizer operation. The initial estimate produced by acquisitionmay differ sufficiently from optimal timing so that performance wouldbenefit from adjustment. The transmit and receive symbol timing clocksmay differ by about 5 ppm, resulting in drift of about 0.1 μS per frame(per a symbol every 8 seconds). This drift must be compensated for. Inpractice, individual independently-delayed signal paths will randomlyrise and diminish in average strength, resulting in situations thatwould be best catered for by different symbol timing. Optimal symboltiming depends on an ability to track these changing situations.

The operation of the symbol timing control is as follows. The approachhas similarities to the early-late gating schemes frequently employed indirect-sequence spread spectrum receivers. As each burst is received, ameasure of the error between the expected preamble and the actualreceived preamble is generated. In addition, in alternating frames,similar measures are made on time advanced and retarded versions of thesame input samples. If no timing adjustment is necessary, the errorgenerated with the existing timing should be less (on average) thaneither of the others. Adjustments are made when this is not the case orthere is a consistent disparity between the advanced and retarded errorestimates. This process is simply a search for bit timing that minimizesthe error statistic.

The control loop used includes an estimator of any consistent change intiming, corresponding to drift with respect to the transmitter. Drift inthe order of 10 ppm can be compensated for by this loop.

This search for a minimum may be hampered by the possible presence of alocal (non-global) minimum. In fact, for this statistic the presence oftwo minima is common. The approach taken to resolve this conflict is asfollows. The more advanced minimum is presumed to be the preferredsampling time. Multiple minima typically arise when there is a smalllevel of delay spread, i.e., less than about 10 μS. Under suchconditions the ratio of magnitudes of the estimated paths in the(symbol-spaced) channel impulse response differs significantly in theregion of the more advanced minimum from that in the more retarded case.Thus, the ratio of tap magnitudes provides a statistic from which toconclude the appropriateness of a selected minimum.

With reference to FIGS. 4A and 4B, they show flow diagrams illustratingthe processing performed by the equalizer 21 to implement bit timingcontrol. Inputs (ox 80) include the on-time and time-offset samples(z(n) and z offset (n)), and a flag to indicate the direction of theoffset. The on-time samples are fed into the equalizer 21 just as theyare during normal training 83. Similarly, the time offset samples arefed to the equalizer 21 (box 84). In both cases, the branch metrics (onthe known correct paths) are accumulated over the latter symbols toprovide measures (ERROR_(cum) and ERROR OFFSET_(cum)) of the degree towhich the samples match expectations.

In a separate process the magnitudes of each of two taps estimated andthe channel impulse response at the end of the training process arecalculated (box 85). Averaging the ratio of these taps over a number offrames (boxes 86-89) permits a judgment to be made as to whether the bittiming has selected an inappropriate local minimum. If a threshold (box90) is reached, then bit timing will be advanced by a full symbol time(box 91). Taking account of the relative time at which samples weretaken (box 92), the ERROR_(cum) and ERROR OFFSET_(cum) measures arecombined to generate a noisy estimate of an appropriate timingadjustment (boxes 93, 94). This estimate is then filtered (box 95) togenerate an actual timing offset adjustment. To compensate forconsistent drift, an additional term "drift₋₋ est" monitors andcompensates for this effect.

Referring back to FIG. 1, the ratio/SNR estimator 40 estimates thecomplex estimated channel impulse response tap values C₀ (k), C₁(k.sub.). Note that the estimated channel impulse response tap values C₀(k), C₁ (k) are generated by the ratio/SNR estimator 40 in the same waythey are generated by the channel impulse response estimator 32, exceptthat they are generated only in response to the preamble for each timeslot, which is a known sequence of symbols. At the same time, signal tonoise ratio (SNR) is estimated by the ratio/SNR estimator 40 by findingthe ratio of signal power (PWR) to mean-square-error (MSE) during thesync. pattern (under the assumption that the transmission channelconsisted of two symbol-spaced paths). That is, where PWR and MSE couldbe calculated as follows: ##EQU3## where; N_(s) is the number of samplesin the `sync` pattern (14 for IS-42) sync(1) through sync N_(s) are the(complex) elements of the `sync` pattern; Z(k) are the complex samplesstored at the output of the matched filter. Note that "noise" in thiscontext can consist of either background noise (EG thermal) oruncorrelated interference (such as a co-channel signal intended foranother cellular transceiver in another cell).

Note that C₀ typically represents a dominant path, and C₁ typicallyrepresents secondary paths.

The ratio/SNR estimator 40 determines the ratio of the dominant path'spower amplitude C₀ to the secondary path's power amplitude C₁. Thisratio is referenced to herein as the path ratio, and is indicative ofthe amount of delay spread in the radio frequency signal. If the ratioof power amplitudes falls below approximately a fixed threshold of 22 dB(i.e., 22 dB±1 dB), then the threshold decision controller 42 activates(or selects) the equalizer 21 by controlling the switching devices 41,49, 48, 46 to assume their second states, wherein the common poles arecoupled to the second poles. Similarly, if the ratio of power amplitudesrises above approximately a fixed threshold of 22 dB (i.e., 22 dB±1 dB),then the threshold decision controller 42 deactivates (or deselects) theequalizer 21 by controlling the switching devices 41, 49, 48, 46 toassume their first states, wherein the common poles is coupled to thefirst poles.

Thus, the equalizer 21 is activated in response to the ratio of poweramplitudes falling below about 22 dB, which is indicative of increasedfading or delay spread in the received cellular telephone radiofrequency signal. In addition, the equalizer 21 is deactivated inresponse to the ratio of power amplitudes rising above about 22 dB,which is indicative of decreased fading or delay spread in the receivedcellular telephone radio frequency signal.

Referring next to FIG. 5, a graph is shown of an exemplary variablethreshold for activation and deactivation (selection or deselection) ofthe equalizer 21. As described above, the graph illustrates that theequalizer 21 is activated (and the low complexity voice processor 39 isalso activated) whenever the ratio of power amplitudes C₀, C₁ (or ratioof channel impulse response tap values) falls below about 22 dB. Themagnitude of this 22 dB threshold 500 is illustrated along the verticalor (ordinate) axis 502. However a third factor, the signal to noiseratio (SNR), is shown as also affecting this threshold. The SNR is shownon a horizontal (or abscissa) axis 504.

In the event the channel consists of only a single path, i.e., thecellular telephone radio frequency signal is not scattered into multiplepaths, the magnitude of the secondary paths' estimated power amplitudeC₁ will increase if the SNR is decreased. This occurs even though thereare in fact no significant secondary paths. Consequently, the ratio ofpower amplitudes C₀, C₁, will decrease below the 22 dB fixed threshold500, even though the channel consists of only the single path. Thus, inaccordance with the present embodiment, the threshold 506 is lowered inresponse to low SNR conditions, i.e., as a function of low SNR. As shownin FIG. 5 the threshold 506 is lowered in response to a decreased SNRafter the SNR reaches about 11 dB. As the SNR approaches 0 dB, thethreshold 506 is lowered to approximately 11 dB, about half of the 22 dBfixed threshold 500.

Thus, the threshold 500, 506 utilized to determine whether or not toactivate the equalizer 21 varies as a function of the SNR at low SNRlevels. As a result, this embodiment is able to accurately determinewhether to activate or deactivate the equalizer 21 even in high noiseconditions.

Furthermore, applying about 1 dB of hysteresis between actual thresholdsensures that switching between modes (activated and deactivated) is nottoo erratic (without significantly impacting the location of thethresholds).

Finally, note that the complex power amplitude C₀, C₁ (channel impulseresponse tap values) can be estimated by averaging over a number ofreceived frames so as to avoid excessive and/or erroneous activation anddeactivation of the equalizer 21. Such estimates are shown below:

    C.sub.0-- power.sub.-- est=(1-Kc).C.sub.0-- power.sub.-- est+Kc.|C.sub.0 |.sup.2

    C.sub.1-- power.sub.-- est=(1-Kc).C.sub.1-- power.sub.-- est+Kc.|C.sub.1 |.sup.2

The signal power (PWR) and the mean-squared-error (MSE) can also beestimated over several received frames:

    PWR.sub.-- est=(1-Kp).PWR.sub.-- est+Kp.PWR

    MSE.sub.-- est=(1-Km).MSE.sub.-- est+Km.MSE

where the constants Kc, Kp, and Km determine the time-constant of theindividual first-order averagers; and C₀₋₋ power₋₋ est, C₁₋₋ power₋₋est, PWR₋₋ est, and MSE₋₋ est are the filtered estimates of each neededquantity.

Thus the present invention provides for the selection of a receiver(i.e., equalized or not equalized) based on the threshold delay spread,which is a function of the signal-to-noise ratio of an incomingtelephone signal.

While the invention herein disclosed has been described by means ofspecific embodiments and applications thereof, numerous modificationsand variations could be made thereto by those skilled in the art withoutdeparting from the scope of the invention set forth in the claims. Forexample, the equalizer 21 need not be a maximum likelihood sequenceestimation based equalizer, but may be any of a number of known types ofequalizers, e.g., linear equalizers and decision feedback equalizers(DFEs). Moreover, the present invention can also be utilized to balanceprocessing between the equalizing receiver 21 and non-equalizingreceiver 18, and other signal processors or processing functions of thedigital signal processor. Such other processors or other processingfunctions can be balanced with the equalizing receiver 21 andnon-equalizing receiver 18 instead of, or in addition to, the high andlow complexity voice processors 29, 39 described above. For example, thepresent invention could be utilized to balance processing between theequalizing receiver 21 and non-equalizing receiver 18, and high and lowcomplexity processors for data decoding during data transfers in asystem that transmits digital data instead of compensated voice. Suchsystems are planned for future application of IS-54.

What is claimed is:
 1. In a digital communications system for receivinga radio frequency signal transmitted over a transmission channel and fortransmitting a transmitter output signal to the transmission channel, amethod of selecting and deselecting an equalizer, and for balancingprocessing in response to such selection, the method including:receivingthe radio frequency signal from the transmission channel into a receivepath; determining whether a preselected characteristic of thetransmission channel exceeds a preselected threshold; selecting theequalizer in the event the preselected characteristic of thetransmission channel exceeds the preselected threshold; deselecting theequalizer in the event the preselected characteristic of thetransmission channel does not exceed the preselected threshold;selecting a high complexity processor in the event the equalizer isdeselected; selecting a low complexity processor in the event theequalizer is selected; generating the transmitter output signal usingthe low complexity processor in the event the low complexity processoris selected; and generating the transmitter output signal using the highcomplexity processor in the event the high complexity processor isselected.
 2. The method of claim 1 further including:estimating anestimated delay spread in the radio frequency signal; determining athreshold delay spread, wherein said selecting of said equalizerincludes selecting said equalizer in the event the estimated delayspread exceeds the threshold delay spread, and wherein said deselectingof said equalizer includes deselecting said equalizer in the event theestimated delay spread does not exceed the threshold delay spread;generating a receiver output signal in response to an equalizer signalgenerated by the equalizer in the event the equalizer is selected, theequalizer receiving the radio frequency signal from the receive path,and generating the equalizer signal in response thereto; and generatingthe receiver output signal in response to the radio frequency signalfrom the receive path in the event the equalizer is deselected.
 3. Themethod of claim 2 wherein the selecting of said high complexityprocessor includes selecting a high complexity voice processor in theevent the estimated delay spread does not exceed the threshold delayspread, and the selecting of said low complexity processor includesselecting a low complexity voice processor in the event the estimateddelay spread exceeds the threshold delay spread.
 4. The method of claim3 wherein the receiving of said radio frequency signal includesdigitizing the radio frequency signal so as to generate samples, andwherein the generating of said receiver output signal includes:storingsequentially a plurality of said samples, the plurality of said samplesbeing received during a time slot; estimating a sample within theplurality of said samples at which a data value estimate error is mostprobable; processing the plurality of said samples, having been stored,starting with a first received sample in the time slot and proceeding ina forward direction with respect to the sequence in which the sampleswere stored, beyond the estimated sample within the plurality of saidsamples, at which the data value estimate error is most probable, usinga maximum likelihood sequence estimation procedure to generate estimatesof the values of the samples; processing the plurality of said samples,having been stored, starting with a last received sample in the timeslot and proceeding in a reverse direction with respect to the sequencein which the samples were stored, beyond the estimated sample within theplurality of said samples at which the data value estimate error is mostprobable, using the maximum likelihood sequence estimation procedure togenerate estimates of the values of the samples; and processing theestimates of the values of the samples to generate enhanced estimates ofthe values of the samples.
 5. The method of claim 4 wherein the step ofprocessing the estimates comprises generating transmission channelimpulse response estimates for use in generating enhanced estimates. 6.The method of claim 5 wherein said estimating of said estimated delayspread includes determining a ratio of said transmission channel impulseresponse estimates.
 7. The method of claim 6 wherein the step ofgenerating said transmission channel impulse response estimatescomprises using variable tap coefficients that are determined byestimating tap settings for said transmission channel impulse responseestimates by minimizing the square of the difference between actualreceived samples and those synthesized by passing known transmittedsignals through the transmission channel, and wherein the generating isdone in an iterative manner by combining previous transmission channelimpulse response estimates with new estimates thereof based on recentestimates, and by varying the ratio of the contributions from theprevious and new estimates as a function of location within said timeslot.
 8. The method of claim 2 wherein said determining of saidthreshold delay spread includes:determining said threshold delay spreadas a function of a signal to noise ratio of said radio frequency signal.9. The method of claim 8 wherein said determining of said thresholddelay spread includes:determining said threshold delay spread as afunction of said signal to noise ratio, wherein said threshold delayspread is substantially constant when said signal to noise ratio isgreater than a prescribed signal to noise ratio level and said thresholddelay spread decreases as a function of decreasing signal to noise ratiowhen said signal to noise ratio is less than the prescribed signal tonoise ratio level.
 10. The method of claim 2 wherein the receiving ofsaid radio frequency signal includes digitizing the radio frequencysignal so as to generate samples, and said generating of said receiveroutput signal includes:storing sequentially a plurality of said samples,the plurality of said samples being received during a time slot;estimating a sample within the plurality of said samples at which a datavalue estimate error is most probable; processing the plurality of saidsamples, having been stored, starting with a first received sample inthe time slot and proceeding in a forward direction with respect to thesequence in which the samples were stored, beyond the estimated samplewithin the plurality of said samples at which the data value estimateerror is most probable, using a maximum likelihood sequence estimationprocedure to generate estimates of the values of the samples; processingthe plurality of said samples, having been stored, starting with a lastreceived sample in the time slot and proceeding in a reverse directionwith respect to the sequence in which the samples were stored, beyondthe estimated sample within the plurality of said samples at which thedata value estimate error is most probable, using the maximum likelihoodsequence estimation procedure to generate estimates of the values of thesamples; and processing the estimates of the values of the samples togenerate enhanced estimates of the values of the samples.
 11. A digitalcommunications system for receiving a radio frequency signal transmittedover a transmission channel and for transmitting a transmitter signal tothe transmission channel, said digital communication receiverincluding:a radio receiver that receives the radio frequency signal fromthe transmission channel into a receive path; an analysis circuitcoupled to the radio receiver, the analysis circuit estimating apreselected characteristic of the transmission channel, and determininga preselected threshold; an equalizer coupled to the radio receiver, theequalizer equalizing the radio frequency signal and generating anequalizer signal in response thereto; a first switching device coupledto the analysis circuit, the equalizer and the radio receiver, the firstswitching device selecting the equalizer in response to the analysiscircuit in the event the preselected characteristic exceeds thepreselected threshold, and deselecting the equalizer in the event thepreselected characteristic does not exceed the preselected threshold; areceiver output coupled to the first switching device, the receiveroutput generating a receiver output signal in response to the equalizersignal from the equalizer in the event the equalizer is selected, thereceiver output generating the receiver output signal in response to theradio frequency signal from the receive path in the event the equalizeris deselected; a radio transmitter that transmits the transmitter signalfrom a transmission path into the transmission channel; a highcomplexity processor coupled to the radio transmitter; a low complexityprocessor coupled to the radio transmitter; a second switching devicecoupled to the high complexity processor and the low complexityprocessor, and receiving a baseband signal, the baseband signal beingcoupled by the second switching device to the high complexity voiceprocessor in the event the preselected characteristic does not exceedthe preselected threshold, and being coupled to the low complexityprocessor in the event the preselected characteristic exceeds thepreselected threshold.
 12. The digital communications receiver of claim11 wherein said preselected characteristic is delay spread, and saidpreselected threshold is a threshold delay spread.
 13. The digitalcommunications receiver of claim 12 wherein the high complexityprocessor includes a high complexity voice processor and wherein the lowcomplexity processor includes a low complexity voice processor.
 14. Thedigital communications receiver of claim 12 wherein the digitalcommunications system includes:a digitizer coupled between the receiver,and the analysis circuit and equalizer, the digitizer digitizing theradio frequency signal so as to generate samples; a memory devicecoupled to the digitizer, the memory device sequentially storing aplurality of said samples, the plurality of said samples being receivedduring a time slot; said analysis circuit estimating a sample within theplurality of said samples at which a data value estimate error is mostprobable; said analysis circuit processing the plurality of saidsamples, having been stored, starting with a first received sample inthe time slot and proceeding, in a forward direction with respect to thesequence in which the samples were stored, beyond the estimated samplewithin the plurality of said samples at which the data value estimateerror is most probable, using a maximum likelihood sequence estimationprocedure to generate estimates of the values of the samples; saidanalysis circuit processing the plurality of said samples, having beenstored, starting with a last received sample in the time slot andproceeding, in a reverse direction with respect to the sequence in whichthe samples were stored, beyond the estimated sample within theplurality of said samples at which the data value estimate error is mostprobable, using the maximum likelihood sequence estimation procedure togenerate estimates of the values of the samples; and said analysiscircuit processing the estimates of the values of the samples togenerate enhanced estimates of the values of the samples.
 15. Thedigital communications receiver of claim 14 wherein said analysiscircuit generates transmission channel impulse response estimates foruse in generating enhanced estimates.
 16. The digital communicationsreceiver of claim 15 wherein said analysis circuit determines a ratio ofsaid transmission channel impulse response estimates in order toestimate said estimated delay spread.
 17. The digital communicationsreceiver of claim 16 wherein said analysis circuit generates the channelimpulse response estimates using variable tap coefficients that aredetermined by said analysis circuit estimating tap settings for saidtransmission channel impulse response estimates by minimizing the squareof the difference between actual received samples and those synthesizedby passing known transmitted signals through a transmission channel, andwherein the processing by said analysis circuit is done in an iterativemanner by combining previous transmission channel impulse responseestimates with new estimates thereof based on recent estimates, and bysaid analysis circuit varying the ratio of the contributions from theprevious and new estimates as a function of location within said timeslot.
 18. The digital communications receiver of claim 12 wherein saidanalysis circuit determines said threshold delay spread as a function ofa signal to noise ratio of said receive signal.
 19. The digitalcommunications receiver of claim 18 wherein said threshold delay spreadis substantially constant when said signal to noise ratio is greaterthan a prescribed signal to noise ratio level and said threshold delayspread decreases as a function of decreasing signal to noise ratio whensaid signal to noise ratio is less than the prescribed signal to noiseratio level.